Electronic power switches have various applications, including the control of light sources such as vehicle headlamps and the like. The current market trend is to use high efficiency light sources such as light emitting diodes (LEDs), which require a very accurate current sense within the power switch/supply. However, in middle to low cost vehicles conventional bulbs are still used within the headlamps, which require a low ON-resistance power switch. Consequently, market demands require power switches that are capable of supporting both types of light sources, and thus that are able to combine a low ON-resistance power switch for heavy loads like conventional bulbs and also have very accurate current sense for light load such as LEDs.
“Smart” power switching circuits, such as for example Freescale's “eXtreme switch” devices, may be used for driving different types of loads such as bulb-lamps or DC-motors. Smart power switching circuits can be configured to perform additional functions such as, for example, protecting the devices against short-circuits, protecting power-MOS elements against over-temperature, providing special and adjustable over-current protection profile required for different bulbs, sensing the current flow at any given time and providing the sense current through a current sense terminal (CSNS), load diagnostics such as open-load detection, load control which may be adapted to requirements by means of pulse width modulation (PWM), suppressing electromagnetic interference during the process of switching, etc.
FIG. 1 illustrates a simplified circuit diagram of an example of a smart power switching module 100. The smart power switching module 100 comprises a power switching device 110 operably coupled between a battery supply 102 and an output 104 of the power switching module 100. The power switching device 110 is controllable via a gate signal 114 to allow a load current (I_LOAD) 112 to flow there through, from the battery supply 102 to the output 104 of the power switching module 100.
The power switching module 100 further comprises current sense component for the load current (I_LOAD) 112. The current sense component comprises a differential or error amplifier 130 comprising a first (inverting) input operably coupled to a source node 115 of the power switching device 110. In this manner, the first (inverse) input of the differential amplifier 130 is arranged to receive a voltage signal representative of the voltage level of the battery supply 102 less the potential difference across the power switching device 110.
The differential amplifier 130 further comprises a second (non-inverting) input arranged to receive a current sense feedback signal generated by a current sense feedback component of the current sense component. The current sense feedback component comprises a sense switching device 120 operably coupled between the battery supply 102 and a ground plane 106. The sense switching device 120 is controllable by the same gate signal 114 as the power switching device 110. In this manner, the current flow through the sense switching device 120 is representative of the current flow through the power switching device 110. A source node 125 of the sense switching device 120 is operably coupled to the second (non-inverting) input of the differential amplifier 130. In this manner, the second (non-inverting) input of the differential amplifier 130 is arranged to receive a voltage signal representative of the voltage level of the battery supply 102 less the potential difference across the sense switching device 120.
A feedback transistor 140 is operably coupled between the source node 125 of the sense switching device 120 and the ground plane 106. A gate of the feedback transistor 140 is operably coupled to a (positive) output of the differential amplifier 130. In this manner, the sense switching device 120 and the feedback transistor 140 are operably coupled in series, with the sense switching device 120 being controllable via the gate signal 114 and the feedback transistor 140 being controllable via the output of the differential amplifier 130. The differential amplifier 130 is arranged to control the current I_SENSE 122 through the feedback transistor 140 such that substantially equal voltage potentials are maintained at its inputs, and thus at the source nodes 115, 125 of the power switching device 110 and sense switching device 120.
Significantly, by maintaining substantially equal voltage potentials at the source nodes 115, 125 of the power switching device 110 and sense switching device 120, and because the sense switching device 120 is controlled by the same gate signal 114 as the power switching device 110, the current I_SENSE 122 is proportional to the load current (I_LOAD) 112. The feedback transistor 140 effectively acts as a voltage to current converter, converting the voltage signal output by the differential amplifier 130 into the current I_SENSE 122. As such, the voltage signal output by the differential amplifier 130 may be considered as being representative of the load current (I_LOAD) 112.
The output of the error amplifier 130 is further used to control an input stage of a current mirror arrangement, whereby the output of the error amplifier 130 is provided to a gate of a voltage-to-current converter transistor 150, which converts the output signal of the error amplifier 130 into an intermediate current I_INT 152 representative of the current I_SENSE 122. A current-to-voltage converter transistor 160, operably coupled in series with the voltage-to-current converter transistor 150, converts the intermediate current I_INT 152 to a voltage signal 162 at its gate terminal. A further voltage-to-current converter transistor 170, which has its gate terminal operably coupled to the gate terminal of the current-to-voltage converter transistor 160, converts the voltage signal 162 into an output sense current I_CSNS 172 which is output through a current sense terminal (CSNS) 180.
The output sense current I_CSNS 172 in the illustrated power switching circuit 100 may be calculated as:I_CSNS=(M/N)*((I_LOAD/RATIO)+(Vos/(RDSON*RATIO)))  [Equation 1]
where:                “RATIO” refers to the electrical ratio between the power and sense transistor devices 110, 120 (“RATIO” is defined as current through the main power switching transistor 110 divided by current through the sense switching device 120 when voltages on their source nodes 115, 125 are equal);        Vos refers to the voltage offset of the error amplifier 130;        RDSON refers to the Drain-to-Source ON resistance of the main power switching device 110; and        the ratio (M/N) refers to current gain for the current mirror circuit (output sense current I_CSNS 172 through the voltage-to-current converter transistor 170 divided by the current I_SENSE 122 through the transistor 140) formed by the two common gate transistor structures of transistors 140 and 150 (N:1) and transistors 160 and 170 (1:M).        
If transistors 140, 150, 160, 170 are such that M=N (e.g. where transistors 140, 150, 160, 170 are all of equal size) the ratio M/N=1 and the term (M/N) may be omitted from Equation 1.
The relationship between the load current I_LOAD 112 and the output sense current I_CSNS 172 is often represented by a current sense ratio (CSR) parameter, where (assuming M/N=1):CSR=I_CSNS/I_LOAD=(1/RATIO)*(1+Vos/(I_LOAD*RDSON))  [Equation 2]
From Equation 2, it can be seen that, since RDSON is substantially constant, the primary contributing factors to CSR inaccuracy for a given I_LOAD are the electrical ratio between main and sense power die devices 110, 120 (RATIO) and the offset of the error amplifier 130 (Vos).
FIG. 2 illustrates a simplified example of the influence of the RATIO and the Vos on the inaccuracy of the CSR. An example of a typical ideal CSR is illustrated at 200. An example of typical CSR inaccuracy caused by variations in the electrical ratio between main and sense power die devices 110, 120 (RATIO), for example due to part-to-part variations, temperature, load current, etc., is illustrated at 210. An example of typical CSR inaccuracy caused by a combination of the electrical ratio between main and sense power die devices 110, 120 (RATIO) and the offset of the error amplifier 130 (Vos) is illustrated at 220. At high load currents, such as illustrated at 230, the primary contributor to CSR inaccuracy is the electrical ratio between main and sense power die devices 110, 120 (RATIO). However, at low load currents the offset of the error amplifier 130 (Vos) becomes the more significant contributor to CSR inaccuracy, which has a significant impact on the accuracy of the current sense component for light loads such as LEDs.